Driving circuit for vibration-type actuator

ABSTRACT

An apparatus includes a vibration-type actuator and a driving circuit. The vibration-type actuator includes a vibration member and a moving member, wherein the vibration member includes an electro-mechanical energy conversion element and the moving member is configured to move relative to the vibration member. The driving circuit includes a capacitor and an inductor connected in series with the capacitor to the electro-mechanical energy conversion element. In an example, parameters of the driving circuit are set such that, when a series resonance frequency of the inductor and the capacitor is denoted by fs and a resonance frequency of the vibration member is denoted by fm, a condition 0.73·fm&lt;fs&lt;1.2·fm is satisfied.

CROSS REFERENCE TO RELATED APPLICATIONS

The present application is a continuation of U.S. patent applicationSer. No. 12/905,993, filed on Oct. 15, 2010, which claims priority fromJapanese Patent Application No. 2009-265234, filed Nov. 20, 2009, all ofwhich are hereby incorporated by reference herein in their entirety.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a driving circuit configured to drive avibration-type actuator.

2. Description of the Related Art

The vibration-type actuator is a non-electromagnetically drivingactuator configured to generate a high-frequency vibration in anelectro-mechanical energy conversion element such as a piezoelectricelement by applying an alternating voltage to the electro-mechanicalenergy conversion element whereby vibration energy is output in the formof continuous mechanical motion. The vibration-type actuators areclassified into a standing wave type and a traveling wave type accordingto types of generated vibrations.

FIG. 16 illustrates a conventional driving circuit configured to drive avibration-type actuator of the traveling wave type (see Japanese PatentPublication No. 5016277). A vibration member 101 is a combination of apiezoelectric element and an elastic element. The piezoelectric elementis applied with an alternating voltage via driving electrodes 101 a and101 b. An oscillator 601 generates an alternating signal correspondingto a driving frequency. A switching circuit 602 operates such that aswitching element in the switching circuit 602 turns on and off inaccordance with the alternating signal supplied from the oscillator 601thereby generating an alternating voltage. The switching circuit 602 isconnected to a DC voltage source (not shown) such that the alternatingvoltage is generated from a DC voltage supplied from the DC voltagesource.

The actuator shown in FIG. 16 employs a two-phase driving scheme. Inthis scheme, alternating voltages with different phases are providedfrom two parts of the driving circuit. These two parts of the drivingcircuit are similar except that the phase of an input alternatingvoltage is shifted by ±90° by a 90°-phase shifter 603. Therefore, thefollowing explanation is given only for a part 604 that is one of thesetwo parts.

The alternating voltage Vi output from the switching circuit 602 isapplied to a primary coil 401 a of a transformer 401, and stepped up byan amount corresponding to the turn ratio of the secondary coil 401 b tothe primary coil 401 a of the transformer 401. The stepped-upalternating voltage Vo is passed through an inductor 102 connected inseries to the secondary coil 401 b of the transformer 401 to removeharmonic components from the waveform of the alternating voltage Vo. Theresultant alternating voltage Vo is applied to the driving electrode 101a. In the actuator disclosed in Japanese Patent Publication No. 5016277,a capacitor 103 is connected to the primary coil 401 a of thetransformer 401 such that series resonance occurs between the capacitor103 and the primary coil 401 a whereby the frequency characteristic ofthe alternating voltage Vo has a peak. Note that the series resonancefrequency of the series of the capacitor 103 and the primary coil 401 aof the transformer 401 is set to be equal to the resonance frequency ofthe vibration member 101. This configuration makes it possible to adjustthe alternating voltage Vo by controlling the driving frequency evenwhen a change occurs in the resonance frequency of the vibration member101, whereby it is possible to reduce the power consumption.

SUMMARY OF THE INVENTION

In the conventional driving circuit for the traveling-wave vibrationactuator, a great change occurs in the alternating voltage Vo applied tothe vibration member 101 in a frequency range from a starting frequencyto an operation frequency corresponding to a specified number ofrotations, i.e., the frequency characteristic of the alternating voltageVo has a steep gradient close to the resonance frequency of thevibration member 101. This results in a change in voltage amplitude,which causes degradation in responsiveness to a driving speed, which inturn causes degradation in controllability. In view of the above, thepresent invention provides a driving circuit having a small change inoutput voltage over a full driving frequency range from a startingfrequency to an operation frequency.

In an aspect of the present invention, there is provided and apparatusthat includes a vibration-type actuator having a vibration member and amoving member, wherein the vibration member includes anelectro-mechanical energy conversion element and the moving member isconfigured to move relative to the vibration member, and includes adriving circuit having a capacitor and an inductor connected in serieswith the capacitor to the electro-mechanical energy conversion element,wherein parameters of the driving circuit are set such that, when aseries resonance frequency of the inductor and the capacitor is denotedby fs and a resonance frequency of the vibration member is denoted byfm, a condition 0.73·fm<fs<1.2·fm is satisfied.

In an aspect of the present invention, there is provided an apparatusthat includes a vibration-type actuator having a vibration member and amoving member, wherein the vibration member includes anelectro-mechanical energy conversion element and the moving member isconfigured to move relative to the vibration member, and a drivingcircuit having a transformer and an inductor and a capacitor, whereinthe transformer includes a primary coil and a secondary coil and isconnected in parallel to the electro-mechanical energy conversionelement, wherein an alternating voltage is configured to be applied tothe primary coil, and wherein the inductor and the capacitor are locatedat least one of on a primary side and a secondary side of thetransformer such that the inductor and the capacitor are connected inseries to the electro-mechanical energy conversion element, whereinparameters of the driving circuit are set such that, when a seriesresonance frequency of the inductor and the capacitor is denoted by fsand a resonance frequency of the vibration member is denoted by fm, acondition 0.73·fm<fs<1.2·fm is satisfied.

Thus, the driving circuit according to any aspect of the presentinvention provides an output voltage with a small change over the fullfrequency range from the starting frequency to the operation frequencyand thus provides improved frequency controllability.

Further features of the present invention will become apparent from thefollowing description of exemplary embodiments with reference to theattached drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIGS. 1A and 1B are diagrams illustrating a driving circuit configuredto drive a vibration-type actuator according to an embodiment of thepresent invention, and FIG. 1C is a diagram illustrating a simulatedcharacteristic thereof.

FIG. 2A is a diagram illustrating a comparative example of a drivingcircuit configured to drive a vibration-type actuator according to aconventional technique, and FIG. 2B is a diagram illustrating asimulated characteristic thereof.

FIG. 3A is a diagram illustrating a simulation result in terms of aphase of an alternating voltage Vo, and FIG. 3B is a diagramillustrating a simulation result in terms of a change in alternatingvoltage Vo as a function of a frequency.

FIG. 4 is a diagram illustrating a simulation result in terms of arelative change in phase with respect to a change that occurs in aconventional technique as a function of fs/fm.

FIG. 5 is a diagram illustrating a simulation result in terms of afrequency characteristic of an alternating voltage Vo for a case where aseries resonance frequency fs is lower than a resonance frequency fm ofa vibration member.

FIG. 6 is a diagram illustrating a relationship between inductance of aninductor and capacitance of a capacitor for a plurality of peakfrequencies fe according to an embodiment of the present invention.

FIG. 7 is a diagram illustrating a frequency characteristic of analternating voltage Vo for a case where fe<1.5·fd.

FIG. 8 is a diagram illustrating a driving circuit configured to drive avibration-type actuator according to a modified embodiment of thepresent invention.

FIG. 9A is a diagram illustrating a driving circuit configured to drivea vibration-type actuator according to an embodiment of the presentinvention, and FIG. 9B is a diagram illustrating a simulatedcharacteristic thereof.

FIG. 10A is a diagram illustrating a driving circuit configured to drivea vibration-type actuator according to a modified embodiment of thepresent invention, and FIG. 10B is a diagram illustrating a simulatedcharacteristic thereof.

FIGS. 11A and 11B are diagrams illustrating simulation results in termsof changes of an alternating voltage Vo due to a variation of a load anda variation of an inductor.

FIGS. 12A to 12E are diagrams illustrating driving circuits configuredto drive a vibration-type actuator according to modified embodiments ofthe present invention.

FIG. 13A is a diagram illustrating a comparative example of a drivingcircuit using a transformer configured to drive a vibration-typeactuator according to a conventional technique, and FIG. 13B is adiagram illustrating a simulated characteristic thereof.

FIGS. 14A and 14B are diagrams illustrating simulation results in termsof changes of an alternating voltage Vo due to a variation of a load anda variation of an inductor of a comparative example of a drivingcircuit.

FIG. 15 is a diagram showing a simulation result in terms of a frequencycharacteristic of an alternating voltage Vo that is output according toa condition described in Japanese Patent Publication No. 5016277.

FIG. 16 is a diagram illustrating a driving circuit disclosed inJapanese Patent Publication No. 5016277.

DESCRIPTION OF THE EMBODIMENTS

The driving circuit of the vibration-type actuator according to thepresent invention is described in further detail below with reference toembodiments in conjunction with the accompanying drawings. The drivingcircuit according to the present invention is applicable to avibration-type actuator that is configured as follows. That is, thevibration-type actuator driven by the driving circuit according to thepresent invention includes a vibration member having anelectro-mechanical energy conversion element such as a piezoelectricelement and an elastic element connected to the electro-mechanicalenergy conversion element, and also includes a moving member that isurged into contact with the elastic element and that moves relative tothe vibration member. The electro-mechanical energy conversion elementis applied with a plurality of alternating voltages that are differentin phase such that a vibration wave is generated in the elastic element.The generated vibration wave causes the elastic element to have anelliptic motion at a driving part (in contact with the moving member) inthe elastic element, and this elliptic motion causes the moving memberto move relative to the vibration member.

In embodiments described below, it is assumed by way of example that thedriving circuit includes two parts, i.e., a first-phase part and asecond-phase part such that the piezoelectric element serving as theelectro-mechanical energy conversion element is driven by alternativevoltages with different phases output from the respective parts. In thisconfiguration, the first-phase part and the second-phase part of thedriving circuit are similar except that a phase of an alternatingvoltage input to each part is shifted by ±90° by a 90° phase shifter603, and thus the following explanation is given only for one part(corresponding to the part 604 shown in FIG. 16). Note that the presentinvention is not limited to the two-phase driving scheme, but thepresent invention is also applicable to other types of driving circuitssuch as a driving circuit configured to drive a traveling-wave-typeactuator by alternating voltages with four or more phases, a drivingcircuit configured to drive a standing-wave-type actuator, etc. Anoscillator that generates an alternative signal and a switching circuitare not essential parts of the present invention, and there is noparticular restriction on these parts. Therefore, the followingdescription is given only for a part which, in the driving circuit shownin FIG. 16, receives an alternating voltage Vi and outputs analternating voltage Vo applied to the vibration member 101.

First Embodiment Example in which an Inductor and a Capacitor areConnected in Series to a Vibration Member

Referring to FIGS. 1A to 1C, a driving circuit according to a firstembodiment is described below. FIG. 1A illustrates the driving circuitof the vibration-type actuator according to the first embodiment. Thedriving circuit is configured such that an inductor 102 and a capacitor103 are connected in series to the vibration member 101 (i.e., in seriesto the electro-mechanical energy conversion element). An inductanceelement such as a coil may be used as the inductor 102, and acapacitance element such as a film capacitor may be used as thecapacitor 103. In the present embodiment of the invention, the seriesresonance frequency of the inductor 102 and the capacitor 103 is set tobe substantially equal to the resonance frequency of the vibrationmember 101.

An equivalent circuit of the vibration member 101 is described belowwith reference to FIG. 1B. FIG. 1B illustrates an equivalent circuit ofthe one-phase part of the vibration member 101. The equivalent circuitof the vibration member 101 includes an RLC series circuit correspondingto a mechanically vibrating part (an equivalent coil 301 b withself-inductance Lm, an equivalent capacitor 301 c with capacitance Cm,and an equivalent resistor 301 d with resistance Rm) and a capacitor 301a with an intrinsic capacitance Cd of the vibration member 101. Notethat the capacitor 301 a is connected in parallel with the RLC seriescircuit.

Hereinafter, the series resonance frequency of the inductor 102 and thecapacitor 103 is denoted by fs, and the resonance frequency of thevibration member 101 is denoted by fm. Furthermore, if theself-inductance of the inductor 102 is denoted by L, and the capacitanceof the capacitor 103 is denoted by C, then fs and fm are given asfollows.

fs=1/(2π√{square root over (LC)})  (1-1)

fm=1/(2π√{square root over (LmCm)})  (1-2)

By setting fs to be substantially equal to fm, it becomes possible toobtain a gradual change in frequency characteristic of the alternatingvoltage Vo in a range close to fm.

FIG. 1C shows a simulated alternating voltage Vo for a case where theseries resonance frequency of the inductor 102 and the capacitor 103 isset to be equal to the resonance frequency of the vibration member 101.In this simulation, parameters were set as follows. The self-inductanceL of the inductor 102 was set to 2 mH, the capacitance C of thecapacitor 103 was set to 6.5 nF, the self-inductance Lm of theequivalent coil 301 b was set to 0.1 H, and the capacitance Cm of theequivalent capacitor 301 c was set to 130 pF. In FIG. 1C, a verticalaxis indicates the gain of the amplitude of the alternating voltage Voat the output side relative to the alternating voltage Vi at the inputside. For example, when the gain of the amplitude is equal to 3, if theamplitude of Vi is 100 V, then the amplitude of Vo is 300 V. As can beseen from FIG. 1C, by setting fs to be equal to fm, it is possible toachieve a gradual change in the frequency characteristic of thealternating voltage Vo in a range close to fm. The change in theamplitude of the alternating voltage Vo in the range close to fm iscaused by a change in impedance of the self-inductance Lm and thecapacitance Cm of the mechanically vibrating part of the vibrationmember 101. In the present embodiment, this problem is reduced bysetting fs to be equal to fm thereby achieving impedance matching withthe impedance of the mechanically vibrating part of the vibration member101 and thus reducing the change in the amplitude of the alternatingvoltage Vo. The reduction in the change in the frequency characteristicof the alternating voltage Vo in the range close to fm leads to areduction in change of the alternating voltage Vo due to a variation ofa load (equivalent resistor 301 d) or the inductor 102. This is becausethe good impedance matching with the mechanically vibrating part of thevibration member 101 is maintained, and thus changes in characteristicof circuit elements do not have a significant influence on the frequencycharacteristic in the frequency range around fm.

Note that the parameters are also set such that an electric resonance ofthe inductor 102 and the capacitor 103 and the capacitor 301 a of thevibration member 101 causes the amplitude of the alternating voltage Voto have a peak at a particular frequency. Hereinafter, the peakfrequency of the alternating voltage Vo is denoted by fe. As can be seenin FIG. 1C, by setting fe to be higher than fm, it is possible to obtaina frequency characteristic with a small voltage change in a frequencyrange from fm to fe regardless of a change in the driving frequency fdof the vibration member 101.

First Comparative Example in which Only Inductor is Connected in Seriesto Vibration Member

Referring to FIG. 2, a discussion is given below for a case where onlythe inductor 102 is connected in series to the vibration member 101.FIG. 2A illustrates a driving circuit in which only an inductor 102 isconnected in serial to the vibration member 101. FIG. 2B shows asimulated frequency characteristic of the alternating voltage Vo for acase in which the circuit shown in FIG. 2A is used. In the simulation,parameters were set such that the electric resonance of the inductor 102and the capacitor 301 a of the vibration member 101 caused the amplitudeof the alternating voltage Vo to have a peak at a particular frequency.More specifically, the parameters were set as follows. Theself-inductance L of the inductor 102 was to be 1.23 mH and theintrinsic capacitance Cd of the capacitor 301 a of the vibration member101 was set to 3.5 nF such that the alternating voltage Vo had a peak ata frequency of 76.707 kHz. Furthermore, in the simulation, the resonancefrequency fm of the vibration member 101 was assumed to be 44.142 kHz.As can be seen from FIG. 2B, the frequency characteristic of thealternating voltage Vo has a great change in voltage in a frequencyrange around fm, which results in degradation in controllability.Another problem is that a steep change occurs in the alternating voltageVo in a range from the resonance frequency fm of the vibration member101 to the peak frequency fe of Vo, and this steep change causes a highvoltage to be output in a high range of the driving frequency.Therefore, circuit elements such as a switching element used in thedriving circuit need to have a high withstand voltage, which causes anincrease in cost. Furthermore, this leads to an increase in thealternating voltage Vo due to a variation of the load (equivalentresistor 301 d) or the inductor 102.

Maximum Allowable Difference Between fs and fm

In the present embodiment of the invention, the series resonancefrequency fs of the inductor 102 and the capacitor 103 connected inseries to the vibration member 101 does not need to be exactly equal tothe resonance frequency fm of the vibration member 101. That is, it ispossible to achieve a gradual change in the frequency characteristic ofthe alternating voltage Vo in a frequency range around fm as long as thedifference between fs and fm is within a particular narrow range,although the smaller the difference is between fs and fm, the betterresult is obtained.

To determine a range of fs in which the advantages of the presentembodiment of the invention are achieved, an investigation is made on aneffect of a change in phase of the alternating voltage Vo in a frequencyrange around the resonance frequency fm of the vibration member 101.FIG. 3A shows a result of simulation in terms of the phase of thealternating voltage Vo where a horizontal axis indicates the frequency,and a change in phase of Vo is shown for a range from 40 kHz to 48 kHzaround a resonance frequency fm set to be equal to 44.142 kHz. Thesimulation was performed for the driving circuit shown in FIG. 1A. Inthe simulation, the series resonance frequency fs of the inductor 102and the capacitor 103 was varied in a range from 0.73 to 1.2 in relativevalue with respect to fm (i.e., fs/fm), and the result is plotted inFIG. 3A. Note that when fs/fm was varied, L and C were adjusted so thatthe peak frequency fe was maintained at 61.798 kHz (=1.4·fm). The reasonwhy the peak frequency fe was maintained at the constant value is that achange in the value of the peak frequency fe causes a great change inthe amplitude of Vo in a frequency range around the resonance frequencyfm of the vibration member 101. For the purpose of comparison, thesimulation was also performed for the circuit according to theconventional technique shown in FIG. 2A, and a result was plotted. Inthe simulation for the circuit shown in FIG. 2A, the self-inductance Lof the inductor 102 was set to 1.97 mH, and the peak frequency fe of thealternating voltage Vo was set to be 61.798 kHz (=1.4·fm).

From FIG. 3A, it can be seen that the conventional circuit configurationhas a great phase delay. The maximum phase delay almost reaches 60°. Incontrast, when fs was set such that fs/fm=1, Vo had substantially nophase change. When fs/fm=1, the self-inductance L of the inductor 102was 4.17 mH and the capacitance C of the capacitor 103 was 3.12 nF.Generally, the phase change increases in a negative direction withdecreasing fs/fm<1, while the phase change increases in a positivedirection with increasing fs/fm>1.

A simulation was also performed in terms of a dependency of thealternating voltage Vo on the frequency to detect a relationship betweenthe phase change of the alternating voltage Vo shown in FIG. 3A and achange of the amplitude of the alternating voltage Vo. The result isshown in FIG. 3B. The simulation was performed under the same conditionas that of FIG. 3A. The change of the alternating voltage Vo wascalculated for values of fs/fm from 0.73 to 1.2, and the result wasplotted. For the purpose of comparison, the change of the alternatingvoltage Vo for the conventional circuit configuration was alsocalculated and the result was plotted. As can be seen, the phase changeshown in FIG. 3B roughly corresponds to the voltage change shown in FIG.3A. That is, the change in the amplitude of Vo increases with increasechange in phase of Vo.

FIG. 4 illustrates a simulation result in terms of a relative change inphase with respect to a change that occurs in the conventionalconfiguration as a function of fs/fm. In FIG. 4, a horizontal axisrepresents fs/fm, i.e., the ratio of fs to the resonance frequency fm ofthe vibration member 101. A vertical axis represents the ratio of thechange in phase to the change in phase that occurs in the conventionalconfiguration. The ratio of the change in phase was calculated asfollows. First, the absolute value of the change in phase of Vo thatoccurs in the conventional configuration was calculated for a frequencyrange from 40 kHz to 48 kHz, and a maximum value was detected.Hereinafter, the detected maximum value is referred to as the maximumphase change in the conventional configuration. Next, for theconfiguration shown in FIG. 1A, the absolute value of the phase changeof Vo was calculated as a function of fs/fm for a frequency range from40 kHz to 48 kHz, and a maximum value was detected. Hereinafter, thedetected maximum value is referred to as the maximum phase changedepending on fs/fm. The ratio of the maximum phase change depending onfs/fm to the maximum phase change in the conventional configuration wasthen calculated, and the result is plotted such that the ratio isrepresented by the vertical axis.

In the present embodiment of the invention, as shown in FIG. 4, when therelative phase change with respect to the phase change of theconventional circuit configuration is defined is smaller than athreshold value set to 0.5, the frequency characteristic of thealternating voltage Vo can be regarded as having a sufficiently smallchange in a frequency range around fm. Such a small change can beachieved when fs/fm is within a range shown below.

0.73·fm<fs<1.2·fm

The above result was obtained when parameters were set as follows. Thepeak frequency fe was set to 61.798 kHz (=1.4·fm), and the intrinsiccapacitance Cd of the capacitor 301 a of the vibration member 101 wasset to 3.5 nF. Note that a similar result is obtained for various valuesof the peak frequency fe and for various values of the intrinsiccapacitance Cd. In the simulation, other parameters were set as follows.The self-inductance Lm of the equivalent coil 301 b of the vibrationmember 101 was set to 0.1 H, the capacitance Cm of the equivalentcapacitor 301 c was set to 130 pF, and the resistance Rm of theequivalent resistor 301 d was set to 1 kΩ.

Thus, by setting fs within the range described above to reduce the phasechange of the alternating voltage Vo to a level less than one half ofthat of the conventional configuration, it also becomes possible toreduce the change in Vo to a level less than one of that of theconventional configuration. That is, even when fs is not exactly equalto fm, if fs and fm satisfy the above-described relationship, it ispossible to reduce the change in frequency characteristic of thealternating voltage Vo in the range around fm compared with theconventional circuit configuration. Thus, it is possible to achieve astable control characteristic due to a synergy effect of the reductionin the change in the alternating voltage Vo and the improvement in phasedelay.

FIG. 5 shows a result of simulation in terms of the frequencycharacteristic of the alternating voltage Vo for a case where therelationship between fs and fm satisfies the above condition (and morespecifically, the series resonance frequency fs is lower than theresonance frequency fm of the vibration member 101). The simulation wasperformed for the circuit configuration shown in FIG. 1A. As shown inFIG. 5, the change in the alternating voltage Vo in a range near theresonance frequency fm is smaller than that shown in FIG. 2B. Note that,in the simulation, the resonance frequency of the vibration member 101was assumed to be 44.142 kHz, and the capacitance of the capacitor 103was intentionally increased by 10% so that the series resonancefrequency fs was set to 0.95·fm, i.e., 42.087 kHz which is smaller byabout 2 kHz than fm. As can be seen from the simulation result, evenwhen fs is not exactly equal to fm, it is possible to reduce the changein the alternating voltage Vo in an frequency range around the resonancefrequency fm.

Determination of Inductance L of Inductor 102 and Capacitance C ofCapacitor 103

Next, a method of determining the capacitance of the capacitor 103 andthe inductance of the inductor 102 is described below. The seriesresonance frequency fs is given by the product of the inductance L ofthe inductor 102 and the capacitance C of the capacitor 103. Therefore,for a given value of fs, there can be an infinite number of combinationsof inductance L and the capacitance C that satisfy the give value of fs.However, if the peak frequency fe of the alternating voltage Vo is firstdetermined, there is only one combination of inductance L andcapacitance C for the given fs.

The peak frequency fe of Vo can be calculated from the inductance L ofthe inductor 102, the capacitance C of the capacitor 103, and theintrinsic capacitance Cd of the capacitor 301 a of the vibration member101 according to equation (1-3) shown below.

$\begin{matrix}{{fe} = {1/\left( {2\; \pi \sqrt{L \cdot \frac{C \cdot {Cd}^{\prime}}{C + {Cd}^{\prime}}}} \right)}} & \left( {1\text{-}3} \right)\end{matrix}$

In practical calculation of the peak frequency fe, the vibration member101 may be regarded as an equivalent capacitor, and its capacitance maybe determined taking into account an effect of the RLC series circuit ofthe mechanically vibrating part. Hereinafter, the resultant capacitanceis denoted by Cd′. For example, when the effect of the RLC seriescircuit of the mechanically vibrating part provides an equivalentcapacitance change of 134 pF, Cd′ may be determined as follows.

Cd′=Cd−134 pF

By determining the value of the peak frequency fe according to equation(1-3), it is possible to determine a relationship between L and C. FIG.6 shows the relationship between the inductance L of the inductor 102and the capacitance C of the capacitor 103 for some values of the peakfrequency fe. A horizontal axis indicates the value of C and a verticalaxis indicates the value of L. In FIG. 6, values of L and C determinedaccording to equation (1-3) are plotted for three values of fe, i.e.,fe=1.4·fm, fe=1.5·fm, and fe=2·fm. In FIG. 6, values of L and C are alsoplotted for a case where the product L and C is given by LmCm, i.e., fora case where the series resonance frequency fs is equal to fm. Asdescribed above, Lm is the self-inductance of the equivalent coil 301 b,and Cm is the capacitance of the equivalent capacitor 301 c. As shown inFIG. 6, for a particular constant value of fe, each curve representinginductance as a function of capacitance intersects the line of LC=LmCmat one point. Each intersection gives optimum values of inductance L andcapacitance C for a case where fs equals fm. For example, if fe=1.4·fm,then L is 4.17 mH and C is 3.12 nF.

The value of fe is discussed in further detail below. In the presentembodiment of the invention, when the driving frequency of the vibrationmember 101 is fd, the peak frequency fe may be set so as to satisfy acondition fe<1.5·fd. The reason for this is described below.

FIG. 7 shows a frequency characteristic of the alternating voltage Vofor a case where fe<1.5·fd. In FIG. 7, 2·fd is a second order harmonicfrequency of the driving frequency fd. It may be better for thealternating voltage Vo to have a waveform similar to a sine wave havingas low harmonic components such as second-order or third-order harmoniccomponents as possible. In practice, the driving waveform of alternatingvoltage Vo has a pulse duty that is not exactly equal to 50%, and thusit may be better to reduce the second-order harmonic component. For theabove reason, by setting the peak frequency fe to a value lower than1.5·fd, it is possible to reduce the amplitude of the second-orderharmonic component of the alternating voltage Vo at the frequency of2·fd to a level smaller than that at the driving frequency fd. Forexample, when the driving frequency fd is 46 kHz, 1.5·fd is 69 kHz. Inthis case, if the inductance L of the inductor 102 is set to be 4 mH,and the capacitance C of the capacitor 103 is set to be 3.25 nF, thenthe peak frequency fe is 61.3 kHz and thus the above-described conditionis satisfied.

Modification of First Embodiment

FIG. 8 illustrates a driving circuit configured to drive avibration-type actuator according to a modification of the firstembodiment of the present invention. In this configuration, an inductor201 for parallel resonance is connected in parallel to the vibrationmember 101. The provision of the inductor 201 for parallel resonancecauses parallel resonance to occur with the capacitor 301 a (theintrinsic capacitance Cd) of the vibration member 101. This makes itpossible to achieve a further reduction in the change in the alternatingvoltage Vo due to a variation of the load (equivalent resistor 301 d) orthe inductor 102. Note that in the present modification, fs may bedetermined from equation (1-1) described above.

Second Embodiment

Next, with reference to FIGS. 9A and 9B, a second embodiment of thepresent invention is described below. The second embodiment is differentfrom the first embodiment described above in that voltage step-up isperformed using a transformer.

FIG. 9A illustrates a driving circuit configured to drive avibration-type actuator according to the second embodiment of thepresent invention. In this configuration of the driving circuit, asecondary coil 401 b of a transformer 401 is connected in parallel tothe vibration member 101 (i.e., the secondary coil 401 b of thetransformer 401 is connected in parallel to the electro-mechanicalenergy conversion element), and a capacitor 103 is connected in seriesto a primary coil 401 a of the transformer 401. A capacitance elementsuch as a film capacitor may be used as the capacitor 103. By reducingthe coupling of the transformer 401, it is possible to increase theleakage inductance of the primary coil 401 a of the transformer 401 andthe leakage inductance of the secondary coil 401 b of the transformer401. These leakage inductances can be employed as the inductor. Theleakage inductances are equivalently represented by an inductor 102 a(leakage inductance of the primary coil 401 a of the transformer 401)and an inductor 102 b (leakage inductance of the secondary coil 401 b ofthe transformer 401). A series resonance circuit is formed by these twoleakage inductances and the capacitor 103. Although the capacitor 103 isconnected to a lower terminal of the primary coil 401 a of thetransformer 401 in the configuration shown in FIG. 9A, the capacitor 103may be connected to an upper terminal of the primary coil 401 a. Theseries resonance frequency of the leakage inductance 102 a of theprimary coil 401 a, the leakage inductance 102 b of the secondary coil401 b, and the capacitor 103 is denoted by fs, and the resonancefrequency of the vibration member 101 is denoted by fm. If the leakageinductance 102 a of the primary coil 401 a of the transformer 401 isdenoted by L1, the leakage inductance 102 b of the secondary coil 401 bof the transformer 401 is denoted by L2, and the turn ratio of thesecondary coil 401 b to the primary coil 401 a is denoted by N, and thecapacitance of the capacitor 103 is denoted by C, then

fs=1/(2π√{square root over ({L ₁+(L ₂ /N ²)}C)})  (2-1)

fm=1/(2π√{square root over (LmCm)})  (2-2)

As described above, Lm and Cm are equivalent circuit constantsassociated with the mechanical vibration of the vibration member 101,where Lm is the self-inductance of the equivalent coil 301 b and Cm isthe capacitance of the equivalent capacitor 301 c.

FIG. 9B is a diagram illustrating a simulation result in terms of afrequency characteristic of the alternating voltage Vo for a case wherethe series resonance frequency fs is set to be equal to the resonancefrequency fm of the vibration member 101. As can be seen from FIG. 9B,as in the first embodiment, by setting fs to be equal to fm, it ispossible to achieve a gradual change in the frequency characteristic ofthe alternating voltage Vo in a range around fm. In the simulation, Lwas set to 20 μH (=L1+L2/N2), C to 650 nF, Lm to 0.1H, Cm to 130 pF, andthe turn ratio N to 10. The reduction in the change in the frequencycharacteristic of the alternating voltage Vo in the range close to fmleads to a reduction in change of the alternating voltage Vo due to avariation of a load (equivalent resistor 301 d) or the inductor 102.Hereinafter, the peak frequency of the alternating voltage Vo is denotedby fe. By setting fe to be higher than fm as shown in FIG. 9B, it ispossible to obtain a frequency characteristic with a small voltagechange in a frequency range from fm to fe regardless of a change in thedriving frequency fd. However, in the case where the transformer isused, the connection of the inductor 102 and the capacitor 103 causesthe alternating voltage Vo to have another peak at a frequency lowerthan the resonance frequency fm of the vibration member 101. That is,the alternating voltage Vo has two peaks at frequencies higher and lowerthan fm. In the present embodiment, fe denotes the higher peak.

As in the first embodiment, the series resonance frequency fs may not beexactly equal to the resonance frequency fm of the vibration member 101.The advantages described above may be achieved by setting the seriesresonance frequency fs within a range around fm so as to satisfy acondition shown below.

0.73·fm<fs<1.2·fm

By setting fs within the above-described range, it is possible toachieve a stable control characteristic due to a synergy effect of thereduction in the change in the alternating voltage Vo and an improvementin phase delay.

In the case where the transformer is used, a coefficient associated withLC in the formula used in calculation of fs varies depending on whetherthe inductor 102 and the capacitor 103 are connected to the primary coilor the secondary coil. Thus, there are four configurations as describedbelow.

(1) L and C are connected to the primary side of the transformer.

(2) L and C are connected to the secondary side of the transformer.

(3) L is connected to the primary coil of the transformer and C isconnected to the secondary coil of the transformer.

(4) C is connected to the primary coil of the transformer and L isconnected to the secondary coil of the transformer.

In the configurations (1) and (2) described above, the coefficient of LCis equal to 1. On the other hand, in the configuration of (3), thecoefficient is N², i.e., the term including LC is given by N²·LC. Thisis because L located on the primary side is equivalent to N²·L on thesecondary side where N is the turn ratio of the transformer. In theconfiguration (4), the coefficient is 1/N², i.e., the term including LCis given by (1/N²)·LC. This is because C located on the primary side isequivalent to (1/N²)·C on the secondary side where N is the turn ratioof the transformer.

Next, FIG. 10 B is a diagram illustrating a simulation result in termsof a frequency characteristic of the alternating voltage Vo for a casewhere the series resonance frequency fs is lower than the resonancefrequency fm of the vibration member 101. This simulation was performedfor a circuit configuration shown in FIG. 10A. The resonance frequencyof the vibration member 101 was assumed to be 44.142 kHz. Thecapacitance of the capacitor 103 was intentionally increased by 10% sothat the series resonance frequency fs was set to 0.95·fm, i.e., 42.087kHz which is smaller by about 2 kHz than fm. As can be seen from FIG.10B, even when fs is not exactly equal to fm, it is possible to reducethe change in the alternating voltage Vo in an frequency range aroundthe resonance frequency fm.

The capacitance of the capacitor 103 and the inductance of the inductor102 may be determined in a similar manner to the first embodiment. Thatis, if the peak frequency fe of the alternating voltage Vo is firstdetermined, then it is possible to uniquely determine a combination ofthe inductance and the capacitance.

As in the first embodiment, when the driving frequency of the vibrationmember 101 is denoted by fd, the peak frequency fe is set such that acondition shown below is satisfied.

fe<1.5·fd

By setting the peak frequency fe so as to satisfy the above condition,it is possible to reduce the second-order harmonic component asdescribed above with reference to FIG. 7. For example, when the drivingfrequency fd is 46 kHz, 1.5·fd is 69 kHz. In this case, in the circuitshown in FIG. 5, if the inductance L of the inductor 102 is set to be 40μH, and the capacitance C of the capacitor 103 is set to be 0.325 μF,then the peak frequency fe is 61.3 kHz and thus the above-describedcondition is satisfied.

First Modification of Second Embodiment

FIG. 10A illustrates a driving circuit configured to drive avibration-type actuator according to a first modification of the secondembodiment of the present invention. In this configuration of thedriving circuit, a secondary coil 401 b of a transformer 401 isconnected in parallel to the vibration member 101, and an inductor 102and a capacitor 103 are connected in series to a primary coil 401 a ofthe transformer 401. Note that the circuit configuration in terms of theinductor 102 and the capacitor 103 is not limited to that shown in FIG.10A as long as the inductor 102 and the capacitor 103 are connected inseries to the primary coil 401 a of the transformer 401. When theinductor 102 is located on the primary side of the transformer 401, theinductance thereof may be as small as 1/N² times the inductance whichwould be necessary when the inductor 102 is located on the secondaryside. Note that N denotes the turn ratio. When the capacitor 103 islocated on the primary side of the transformer 401, the withstandvoltage of the capacitor 103 may be as small as 1/N times the withstandvoltage which would be necessary when the capacitor 103 is located onthe secondary side.

If the inductance of the inductor 102 is denoted by L and thecapacitance of the capacitor 103 is denoted by C, the series resonancefrequency fs is given by equation (2-3) shown below, which is the sameas equation (1-1) described above.

fs=1/(2π√{square root over (LC)})  (2-3)

By setting the series resonance frequency fs determined accordingequation (2-3) so as to be equal to the resonance frequency fm of thevibration member 101, it is possible to achieve a gradual change in thefrequency characteristic of the alternating voltage Vo in a range aroundfm. The reduction in the change in the frequency characteristic of thealternating voltage Vo in the range close to fm leads to a reduction inchange of the alternating voltage Vo due to a variation of a load(equivalent resistor 301 d of the mechanical vibration of the vibrationmember 101) or the inductor 102. FIGS. 11A and 11B illustrate effects ofthe reduction in change in the alternating voltage Vo. FIGS. 11A and 11Bshow simulation result in terms of change in the alternating voltage Vodue to variations of a load and the inductance of the inductor 102 forthe circuit shown in FIG. 10A. More specifically, FIG. 11A illustrates asimulation result in terms of a change in the alternating voltage Vo dueto a variation of the load. In FIG. 11A, to provide a betterunderstanding about the change around the resonance frequency fm of thevibration member 101, the result is shown only for a frequency rangefrom 40 kHz to 50 kHz in a horizontal axis. The calculation wasperformed for three different values of the load, i.e., a referencevalue and the reference value±20%. This variation of the load wasassumed to appear as a change in the equivalent resistance Rm in theequivalent circuit of the vibration member 101. The result shown in FIG.11A indicates that the variations of the load have substantially noeffect on the alternating voltage Vo, thus high controllability can beachieved.

FIG. 11B shows a simulation result in terms of a change in thealternating voltage Vo due to a variation of the inductance of theinductor 102. The calculation was performed for three different valuesof the inductance of the inductor 102, i.e., a reference value and thereference value±20%. The result shown in FIG. 11B indicates thevariation of the inductance of the inductor 102 does not have asignificant influence on the alternating voltage Vo. That is, when thedriving circuit has two or more phases, the variation of the inductanceof the inductor 102 does not have a significant influence, and thus itis possible to reduce unevenness in the travelling wave.

In the present embodiment, the transformer 401 may have leakageinductance. In this case, it is necessary to take into count the effectof the leakage inductance in the calculation of the series resonancefrequency fs substantially equal to the resonance frequency fm of thevibration member 101.

Second Modification of Second Embodiment

FIG. 12A illustrates a driving circuit configured to drive avibration-type actuator according to a second modification of the secondembodiment of the present invention. In this configuration of thedriving circuit, a secondary coil 401 b of a transformer 401 isconnected in parallel to the vibration member 101, a capacitor 103 isconnected in series to a primary coil 401 a of the transformer 401, andan inductor 102 is connected in series to the secondary coil 401 b ofthe transformer 401. When the inductor 102 is located on the secondaryside of the transformer 401, the maximum allowable current of theinductor 102 may be as small as 1/N of the maximum allowable currentthat would be necessary when the inductor 102 is located on the primaryside of the transformer 401. Note that N denotes the turn ratio. If theinductance of the inductor 102 is L, and the capacitance of thecapacitor 103 is C, then the series resonance frequency fs is given byequation (2-4) shown below.

fs=1/(2π√{square root over (LC/N ²)})  (2-4)

By setting the series resonance frequency fs determined accordingequation (2-4) so as to be equal to the resonance frequency fm of thevibration member 101, it is possible to achieve a gradual change in thefrequency characteristic of the alternating voltage Vo in a range aroundfm, and it is also possible to reduce the change in the alternatingvoltage Vo due to the variation of the load (equivalent resistor 301 d)or the inductance of the inductor 102.

Third Modification of Second Embodiment

FIG. 12B illustrates a driving circuit configured to drive avibration-type actuator according to a third modification of the secondembodiment of the present invention. In this configuration of thedriving circuit, a secondary coil 401 b of a transformer 401 isconnected in parallel to the vibration member 101, an inductor 102 isconnected in series to a primary coil 401 a of the transformer 401, anda capacitor 103 is connected in series to the secondary coil 401 b ofthe transformer 401. When the inductor 102 is located on the primaryside of the transformer 401, the inductance thereof may be as small as1/N² of the inductance which would be necessary when the inductor 102 islocated on the secondary side. When the capacitor 103 is located on thesecondary side of the transformer 401, the capacitance thereof may be assmall as 1/N² of the capacitance which would be necessary when thecapacitor 103 is located on the primary side. In this configuration, theseries resonance frequency fs is given by equation (2-5) shown below.

fs=1/(2π√{square root over (LC·N ²)})  (2-5)

The series resonance frequency fs determined according equation (2-5) isset to be equal to the resonance frequency fm of the vibration member101.

Fourth Modification of Second Embodiment

FIG. 12C illustrates a driving circuit configured to drive avibration-type actuator according to a fourth modification of the secondembodiment of the present invention. In this configuration of thedriving circuit, a secondary coil 401 b of a transformer 401 isconnected in parallel to the vibration member 101, and an inductor 102and a capacitor 103 are connected in series to a secondary coil 401 b ofthe transformer 401. When the inductor 102 is located on the secondaryside of the transformer 401, the maximum allowable current of theinductor 102 may be as small as 1/N of the maximum allowable currentthat would be necessary when the inductor 102 is located on the primaryside of the transformer 401. When the capacitor 103 is located on thesecondary side of the transformer 401, the capacitance thereof may be assmall as 1/N² of the capacitance which would be necessary when thecapacitor 103 is located on the primary side. In this configuration, theseries resonance frequency fs is given by equation (2-6) shown below.

fs=1/(2π√{square root over (LC)})  (2-6)

The series resonance frequency fs determined according equation (2-6) isset to be equal to the resonance frequency fm of the vibration member101.

Fifth Modification of Second Embodiment

FIG. 12D illustrates a driving circuit configured to drive avibration-type actuator according to a fifth modification of the secondembodiment of the present invention. In this configuration, a resistor2801 is connected in parallel to a primary coil 401 a of a transformer401. As described above, when a transformer is used as in the secondembodiment and first to fourth modifications thereof, the connection ofthe inductor 102 and the capacitor 103 causes the alternating voltage Voto have two peaks at frequencies higher and lower than fm. In thepresent modification, the provision of the resistor 2801 leads to areduction in the peak at the lower frequency. The reduction in the peakat the lower frequency makes it possible to reduce an influence ofdisturbance in a low frequency range and an influence of a variation ofthe load.

Sixth Modification of Second Embodiment

FIG. 12E illustrates a driving circuit configured to drive avibration-type actuator according to a sixth modification of the secondembodiment of the present invention. In this sixth modification, aresistor 2801 and an inductor 2901 for parallel resonance are connectedin parallel to a primary coil 401 a of a transformer. In this sixthmodification, the inductor 2901 for parallel resonance provides a moreeffective reduction in the peak in the low frequency range. Note that inthe present modification, fs can be determined according to equation(2-5) described above.

Second Comparative Example in which Only an Inductance is Connected inSeries to a Vibration Member

Next, with reference to FIGS. 13A and 13B, a comparative example of aconfiguration is described in which a transformer is used and only aninductor 102 is connected in series to the vibration member 101. FIG.13A illustrates a conventional driving circuit using a transformer fordriving a vibration-type actuator. In this circuit configuration, aninductor 102 is connected in series to a secondary coil of thetransformer 401. FIG. 13B shows a simulation result in terms of afrequency characteristic of an alternating voltage Vo output from thesecondary side of the transformer 401 in the circuit shown in FIG. 13A.In the simulation, the inductance L of the inductor 102 was set to be1.23 mH, the alternating voltage Vo was assumed to have a peak at afrequency of 76.707 kHz, and the resonance frequency of the vibrationmember 101 was assumed to be 44.142 kHz. As shown in FIG. 13B, thefrequency characteristic of the alternating voltage Vo has a greatchange in voltage around fm, which leads to a reduction incontrollability. A steep change occurs in the alternating voltage Vo ina range from the resonance frequency fm of the vibration member 101 tothe peak frequency fe of Vo, and this steep change causes a high voltageto be output in a high range of the driving frequency. Therefore,circuit elements such as a switching element used in the driving circuitneed to have a high withstand voltage, which causes an increase in cost.

FIGS. 14A and 14B show simulation result in terms of change in thealternating voltage Vo due to variations of a load (equivalent resistor301 d) and the inductance of the inductor 102 for the circuit shown inFIG. 13A.

More specifically, FIG. 14A illustrates a simulation result in terms ofa change in the alternating voltage Vo due to a variation of the load.To show more clearly the change in the alternating voltage Vo around theresonance frequency of the vibration member 101, the result is shownonly for a frequency range from 40 kHz to 50 kHz in a horizontal axis.The calculation was performed for three values of the load, i.e., areference value and the reference value±20%, This variation of the loadwas assumed to appear as a change in the equivalent resistance of themechanical vibration in the equivalent circuit of the vibration member101. The result shown in FIG. 14A indicates that the variation of theload causes a great change in the frequency characteristic of thealternating voltage Vo, which results in degradation in controllability.

FIG. 14B shows a simulation result in terms of a change in thealternating voltage Vo due to a variation of the inductance of theinductor 102. It was assumed that a coil was used as the inductor 102and the calculation was performed for three different values of theinductance of the inductor 102, i.e., a reference value and thereference value±20%. As shown in FIG. 14B, the variation of theinductance of the inductor 102 causes a great change in the frequencycharacteristic of the alternating voltage Vo. Therefore, when thedriving circuit has two or more phases, if the inductance of the coil isdifferent among the phases, the amplitude of the alternating voltage Vooutput from the driving circuit becomes different between the phases.That is, alternating voltages Vo with different amplitudes are appliedat the same time to the respective driving electrodes 101 a and 101 b ofthe vibration member 101 shown in FIG. 16, which causes unevenness inthe travelling wave.

Third Comparative Example in which the Series Resonance Frequency ofCapacitor 103 and Primary Coil 401 a of Transformer is Set to be Equalto fm.

FIG. 15 shows a simulation result in terms of a frequency characteristicof an alternating voltage Vo output from a secondary coil of atransformer 401 for a case where the series resonance frequency of acapacitor 103 and a primary coil 401 a of the transformer 401 is set tobe equal to the resonance frequency fm of the vibration member 101. Thesimulation was performed for a circuit configuration shown in FIG. 16.The transformer 401 was assumed to have an ideal coupling (with acoupling coefficient of 1) with no leakage inductance. Furthermore, itwas also assumed that the inductance of the primary coil 401 a of thetransformer 401 was 150 μH and the inductance of the secondary coil 401b of the transformer 401 was 15 mH. The capacitance of the capacitor 103connected to the primary coil 401 a of the transformer 401 was set to86.7 nF, and the inductance of the inductor 102 connected to thesecondary coil 401 b of the transformer 401 was set to 1 mH. Theresonance frequency fm of the vibration member 101 was assumed to be44.142 kHz. Note that the series resonance frequency of the capacitor103 and the inductor 102 is 170.96 kHz which is greatly different fromfm. The peak frequency fe of the alternating voltage Vo was 190.927 kHz.

The simulation result shown in FIG. 15 indicates that the frequencycharacteristic of the alternating voltage Vo has a great voltage changearound fm. The simulation also has revealed that the alternating voltageVo has a great change in phase in a frequency range around fm. That is,Japanese Patent Publication No. 5016277 discloses only the seriesresonance of the capacitor 103 and the primary coil 401 a of thetransformer, but Japanese Patent Publication No. 5016277 does notdisclose the technique to adjust the series resonance frequency of thecapacitor 103 and the inductor 102. Thus, in the technique disclosed inJapanese Patent Publication No. 5016277, a great change can occur in thealternating voltage Vo in a frequency range around fm. Furthermore, thesteep change in the alternating voltage Vo in a range from the resonancefrequency fm of the vibration member 101 to the peak frequency fe of Vocan cause a high voltage to be output in a high range of the drivingfrequency. Therefore, circuit elements such as a switching element usedin the driving circuit need to have a high withstand voltage, whichcauses an increase in cost. Furthermore, this leads to an increase inthe alternating voltage Vo due to a variation of the load (equivalentresistor 301 d) or the inductor 102.

Note that the circuit configuration shown in FIG. 16 may be used in thepresent invention. In this circuit configuration shown in FIG. 16, bysetting fs such that 0.73·fm<fs<1.2·fm, it is possible to achieve agradual change in the frequency characteristic of the alternatingvoltage Vo in a range around fm.

While the present invention has been described with reference toexemplary embodiments, it is to be understood that the invention is notlimited to the disclosed exemplary embodiments. The scope of thefollowing claims is to be accorded the broadest interpretation so as toencompass all such modifications and equivalent structures andfunctions.

What is claimed is:
 1. An apparatus comprising: a vibration-typeactuator having a vibration member and a moving member, wherein thevibration member includes an electro-mechanical energy conversionelement and the moving member is configured to move relative to thevibration member; and a driving circuit having a capacitor and aninductor connected in series with the capacitor to theelectro-mechanical energy conversion element, wherein parameters of thedriving circuit are set such that, when a series resonance frequency ofthe inductor and the capacitor is denoted by fs and a resonancefrequency of the vibration member is denoted by fm, a condition0.73·fm<fs<1.2·fm is satisfied.
 2. The apparatus according to claim 1,wherein, if a peak frequency of an alternating voltage applied to theelectro-mechanical energy conversion element is denoted by fe and adriving frequency of the vibration member is denoted by fd, a conditionfe<1.5·fd is satisfied.
 3. The apparatus according to claim 1, whereinthe vibration member is configured to generate a vibration wave inresponse to an alternating voltage applied to the electro-mechanicalenergy conversion element.
 4. The apparatus according to claim 1,wherein the moving member is configured to move in response to avibration wave relative to the vibration member.
 5. An apparatuscomprising: a vibration-type actuator having a vibration member and amoving member, wherein the vibration member includes anelectro-mechanical energy conversion element and the moving member isconfigured to move relative to the vibration member; and a drivingcircuit having a transformer and an inductor and a capacitor, whereinthe transformer includes a primary coil and a secondary coil and isconnected in parallel to the electro-mechanical energy conversionelement, wherein an alternating voltage is configured to be applied tothe primary coil, and wherein the inductor and the capacitor are locatedat least one of on a primary side and a secondary side of thetransformer such that the inductor and the capacitor are connected inseries to the electro-mechanical energy conversion element, whereinparameters of the driving circuit are set such that, when a seriesresonance frequency of the inductor and the capacitor is denoted by fsand a resonance frequency of the vibration member is denoted by fm, acondition 0.73·fm<fs<1.2·fm is satisfied.
 6. The driving circuitaccording to claim 5, wherein the inductor is a leakage inductance ofthe transformer.
 7. The driving circuit according to claim 5, wherein,if a peak frequency of an alternating voltage applied to theelectro-mechanical energy conversion element is denoted by fe and adriving frequency of the vibration member is denoted by fd, a conditionfe<1.5·fd is satisfied.
 8. The apparatus according to claim 5, whereinthe vibration member is configured to generate a vibration wave inresponse to an alternating voltage applied to the electro-mechanicalenergy conversion element.
 9. The apparatus according to claim 5,wherein the moving member is configured to move in response to avibration wave relative to the vibration member.